1. Field of the Invention
The present invention relates to a switching regulator for producing a pulse-width modulated switching signal used for controlling a switching element connected between an input DC signal and a rectifier so that an AC signal such as a rectangular-wave signal is produced from the input DC signal by turning-on and -off the switching element at timings determined by the pulse-width modulated switching signal and a predetermined output voltage is produced from the rectifier by rectifying the AC signal.
2. Description of the Related Art
FIG. 13 shows a construction of a conventional separately-exited forward switching regulator. In FIG. 1, 1 designates a power source input section for receiving an input voltage V.sub.I and 2 designates a switching element to be turned on and off for obtaining an AC voltage such as a rectangular-wave voltage from the input voltage V.sub.I. A semiconductor element such as a transistor, an FET or an IGBT is used for this switching element. 3 designates an isolation transformer for applying an AC voltage to the next stage circuit and 4 designates a rectification smoothing circuit for rectifying and smoothing the AC voltage outputted from the transformer 3 thereby obtaining an output voltage V.sub.0. The rectification smoothing circuit is structured by diodes 4a and 4b, a choke coil 4c and a capacitor 4d. 5 designates a load to which the output voltage V.sub.0 is supplied.
6 designates a voltage dividing circuit, structured by a resistance element, for dividing the output voltage V.sub.0 and detecting variations V.sub.01, V.sub.02, etc. of V.sub.0. 7 designates a triangular wave signal generating circuit for generating a triangular wave signal 8 of a predetermined frequency. This triangular wave signal generating circuit has a known structure such that the charging and discharging of a capacitor are controlled by a transistor, thereby generating a triangular wave signal 8, 9 designates a comparator including an error amplifier for comparing the output voltage of the voltage dividing circuit 6 with the triangular wave signal 8 and amplifying the difference therebetween thereby outputting a pulse-width modulated control signal. 10 designates a driving circuit for converting the pulse-width modulated control signal into a switching signal 11 which is adapted for driving the switching element 2.
The operation according to the above-described structure will be explained below.
The input voltage V.sub.I received by the power source input section 1 is converted into an AC voltage such as a rectangular wave voltage by turning-on and -off the switching element 2 as controlled by the switching signal 11 obtained from the drive circuit 10. This AC voltage is rectified and smoothed by the rectification smoothing circuit 4 as applied thereto through the transformer 3 and the output voltage V.sub.0 is supplied to the load 5.
The output voltage V.sub.0 is detected by the voltage dividing circuit 6 and is added to the comparator 9, which compares the output of the voltage dividing circuit 6 with the triangular wave signal 8 obtained from the triangular wave signal generating circuit 7. The pulse-width modulated control signal obtained by the comparator is used through the drive circuit 10 as the switching signal 11 to control the turning-on and -off of the switching element 2. With the above-described structure and operation, the output voltage V.sub.0 is controlled to be constant.
FIG. 14 shows a timing chart for explaining the operation of the comparator 9. FIG. 14A shows the relation between the output voltage V.sub.0 and the triangular wave signal 8 when the output voltage V.sub.0 changes to V.sub.01 or V.sub.02. FIG. 14B shows the pulse width of the pulse-width modulated control signal obtained when the output voltage V.sub.0 changes to V.sub.01 or V.sub.02. It is clear from FIG. 14B that the pulse width becomes smaller when the output voltage changes to a higher voltage V.sub.01 and becomes large when the output voltage changes to a lower voltage V.sub.02.
The output voltage V.sub.0 is stabilized by turning-on and -off the switching element 2 according to the pulse-width modulated pulse voltage as described above. It is known that in the ideal circuit there exists the following relation between the turning-on to turning-off ratio (called a time ratio) and the output voltage V.sub.0. EQU V.sub.0 =TON/(TON+TOFF).times.NS/NP.times.V.sub.I ( 1)
where, TON represents the time during which the pulse voltage is at the high level H in FIG. 14, TOFF represents the time during which the pulse voltage is at the low level L in FIG. 14. NP represents the number of turns of the primary winding of the transformer 3 in FIG. 13 and NS represents the number of turns of the secondary winding thereof.
As is clear from the above explanation and expression (1), when the output voltage V.sub.0 has changed by the load or when the input voltage V.sub.I has changed, a constant output voltage V.sub.0 can be obtained by changing the time ratio of TON to TOFF.
In the above-described pulse-width modulation control type of the separately excited forward switching regulator, a triangular wave signal of a constant frequency is used so that when the load is stable and the input voltage is constant, the switching element is turned on and off at a constant frequency. In this case, a conduction noise which is conducted inside the circuit and a radiation noise which is radiated outside the circuit also has respective peaks at specific frequency.
FIG. 15 shows one example of the above case. Conduction noise included in the input voltage is measured by a spectral analyzer. The ordinate indicates the noise voltage level and the abscissa indicates the frequency. In FIG. 15, the noise indicated by A is a noise according to the fundamental wave of the turning-on and -off frequency of the switching element (switching frequency), the noise indicated by B is a noise of higher harmonic thereof, and the noises indicated by C and D are known as noises which are generated by leakage magnetic flux attributable to reverse recovery characteristics of the secondary side rectifier diode or the arrangement in winding of the transformer.
Although FIG. 15 shows an example of the spectral analysis of conduction noise, the radiation noise is also generated in a manner similar to the conduction noise and has the same frequency characteristics as those of the conduction noise.
The above-described noises generate undesired sounds and/or adversely affect various kinds of other electronic devices. In order to restrict the peak value of the noise level, a noise filter circuit or the like made of a capacitor and an inductor has been used conventionally. This method is effective in attenuating the peak level of the generated noise. However, in order to attenuate the peak level sufficiently, it has been necessary to use expensive capacitor and inductor having excellent attenuation characteristics or to connect in multi-stage filter circuits each made of a capacitor and an inductor. Each of these methods has problems in that it is expensive and the space for assembling the circuit becomes larger by the multi-stage connection.
Apart from the above methods, a method is known for dispersing the fundamental frequency of the switching within a certain frequency range so as to prevent the noise peaks from occurring at a specific frequency. This method is disclosed in, for example, Japanese Patent Unexamined Publication JP-A-63-69465.
According to the method disclosed in the above publication, the frequency of the triangular wave signal used for determining the fundamental frequency of the switching is changed at random within a given range so that the fundamental frequency component and higher harmonic component of noise are dispersed within a given frequency range.
The method disclosed in the above publication seems to be effective to restrict the noise peak level. However, when this method is actually applied to the separately excited forward switching regulator shown in FIG. 13, the following problems have occurred.
According to the above publication, the restriction of the noise peak level is achieved by changing the frequency of the triangular wave signal within a range of f.sub.p +1/2fb to f.sub.p -1/2fb, where f.sub.p is a center frequency of the triangular wave signal and fb is a maximum range in variation of the triangular wave signal and the effect of noise dispersion is larger as fb is larger. In order to apply the above method to the separately excited forward switching regulator shown in FIG. 13 the present inventors conducted experimental study under a condition of fp=100 KHz and fb=100 KHz. In other words, the frequency of the triangular wave signal was freely changed within the frequency range of 50 to 150 KHz. In order to carry out the experimental study under the above condition, the transformer 3 of the separately excited forward switching regulator shown in FIG. 13 was first designed.
Usually, when the transformer of the separately excited forward switching regulator is designed, magnetic characteristics of the magnetic material used for the transformer are taken into consideration.
Maximum variation .DELTA.B of the magnetic flux density of the transformer is determined by the input voltage V1 applied to the primary winding of the transformer, a turning-on time (ton) for which the switching element is turned on in one cycle of the switching frequency of the switching element, a number Np of turns of the primary winding of the transformer and a sectional area S of the magnetic core on which the primary winding is wound and represented by the following expression. EQU .DELTA.B=V1.times.ton/(Np.times.S) (2)
The possible maximum value of .DELTA.B is limited by the magnetic material of the transformer, the switching frequency and the ambient temperature. For example, in the case of soft ferrite that is widely used for the transformer of the switching regulator, the range of .DELTA.B is from about 0.2 to 0.4 T (tesla) for the transformer of the separately excited forward switching regulator using one switching element when the frequency is 100 KHz and ambient temperature is 60 degrees.
According to the above conditions, when a transformer having a primary winding of Np=20 turns for use with a switch regulator operating at a condition that the input voltage V1 is 130 volts (DC) and the switching frequency is equal to the fundamental frequency 100 KHz is made of a soft ferrite which is operative with an upper limit of .DELTA.B being 0.3 T, the size (cross-section) S of the transformer is calculated as follows: EQU S=108.3 mm.sup.2.
In the above calculation, the time ton is assumed to be 0.5 times of one period of the switching frequency. It is well known in the art that the upper limit of the time ton is set to 0.5 times of one period of the switching frequency when designing the transformer of the separately excited forward switching regulator.
However, in implementing the method of the above publication by using the transformer having the above-described cross section S, a magnetic saturation occurs when the randomly changing frequency becomes lower than 100 KHz. As is clear from the expression (2), when the switching frequency becomes lower, or when the time ton is increased, .DELTA.B increases.
Since the rate of the increase of .DELTA.B is inversely proportional to the switching frequency, immediately when the switching frequency is changed to the minimum frequency 50 KHz, .DELTA.B becomes 0.6 T which is two times of the value of .DELTA.B taken when the frequency is 100 KHz. Thus, the value of .DELTA.B substantially exceeds the upper limit 0.3 T of .DELTA.B of the transformer material as used, and the magnetic saturation occurs. When the transformer is saturated, heating of the transformer increases rapidly and this heating thermally destroys windings and bobbin portions which constitute the transformer or extremely reduces the life of the electronic parts such as the capacitors and semiconductor elements provided around the transformer. Further, the current peak of the primary winding increases thereby to destroy the electronic parts on the primary side.
To solve the above problems, there is a method of increasing the number of turns of the primary winding of the transformer. This method, however, has a problem that, along with the increase of the number of turns, copper loss produced in the winding increases, the space for the winding (transformer window area) increases, resulting in the necessity of using a large size transformer. There is also a method of increasing the cross section of the transformer. However, in the above-described example, when the switching frequency is 50 KHz, the transformer cross section required for preventing the value of .DELTA.B from exceeding its upper limit is S=216.4 mm.sup.2, which is two times of the value designed for the frequency of 100 KHz. This extremely increases the space of assembly on the substrate and also makes the cost very expensive.
There also arises a problem in the smoothing operation at the secondary side. A triangular wave ripple is superimposed on a DC current flowing through a smoothing choke coil 4c in FIG. 13. The ripple current value is obtained by the following expression. EQU Irip=V.sub.0 /(L.times.f).times.(1-NP/NS.times.V.sub.0 /V.sub.I) (3)
where Irip represents a ripple current, L an inductance of the choke coil and f a switching frequency.
As is clear from the expression (3), when the switching frequency f becomes lower, Irip increases, and when the switching frequency f changes from 100 KHz to 50 KHz in the above example, the ripple current becomes two times of the ripple current at the frequency 100 KHz.
The ripple current component which flows through the choke coil causes a fluctuation component of the output voltage V.sub.0 as multiplied by the impedance component of the smoothing capacitor 4d in FIG. 13.
Accordingly, the increment of the ripple current directly causes the fluctuation component of the output voltage. In order to solve this problem, it is necessary to use a large inductance of the choke coil. For this purpose, it is necessary to use a large size of the choke coil itself, which extremely increases the space for assembly on the substrate and also makes the cost very expensive. Alternatively, there is a method of using low impedance parts, which, however, results in a larger size of each part and high costs similarly to the choke coil case.
As described above, in the separately excited forward switching regulator, when the method disclosed in the above-mentioned publication is implemented, it is possible to expect a noise dispersion effect or an effect of reduction of the noise peak level by increasing the range in variation of the frequency of the triangular wave signal from its center frequency. However, it is practically difficult to implement this method because of increase in loads of the transformer and the smoothing coil and capacitor at the secondary side, each having frequency dependent characteristics. It would be easy to reduce the loads by reducing the range in variation of the frequency. However, this has a possibility of decreasing the noise peak reduction effect obtained by dispersing the noise frequency within a certain frequency range.
As explained above, the range in variation of the frequency of the triangular wave signal from its center frequency is a very significant problem for electronic parts having characteristics dependent on the switching frequency.
There is also a problem in the timing for modulating the frequency of the triangular wave signal. The present inventors carried out experimental study while selecting as the above timing an integer times the period of the center frequency of the triangular wave signal. For example, a timer was used to produce a trigger signal at every ten periods of the frequency of 100 KHz and the triangular wave signal was modulated in response to this trigger signal. As a result of the experimental study, the effect of noise dispersion or the reduction of the noise peak level was observed by about 5 to 10% from the level before the modulation, but sufficient reduction was not obtained yet.
Next, it was tried to set the trigger period at the center frequency, but this was unsuccessful because the triangular wave signal was subjected to hunting. Usually, the triangular wave signal is produced by charging and discharging a capacitor. However, the hunting occurred because it was tried to conduct the modulation during charging or discharging of the capacitor. The interval and the timing of the modulation of the triangular wave signal is also an important factor in the actual implementation, as described above. However, this point is not described in the above publication or other literatures.
The above explains the problems which occur in the case of the separately excited forward switching regulator. Similar problems also occur in the case of the separately excited chopper type switching regulator. The separately excited chopper type has no isolation transformer in its structure unlike the separately excited forward type. However, the problems that occur in the electronics parts having frequency dependency, such as the smoothing choke and the capacitor are the same as the problems in the separately excited forward type. Further, the separately excited flyback switching regulator is different in smoothing operation at the secondary side from the separately excited forward switching regulator. However, the problems that occur in the isolation transformer are the same for both types of the switching regulators.
As described above, an attempt to reduce the peak values of the conduction noise and radiation noise by changing at random the frequency of the triangular wave signal has a problem that the transformer and the secondary side smoothing choke coil and capacitor having frequency dependent characteristics are subjected to excessive loads when the range in modulation of the frequency increases.
Further, in the actual implementation, it is necessary to produce a random signal used for randomly modulating the frequency. As a generally known method, there is a method of obtaining a random signal by a DA conversion after building up a logic according to the Monte Carlo method or the like by a circuit made of digital IC's. These methods, however, require at least three IC's including an IC for logic, an IC for a clock and an IC for DA conversion, requiring a complex circuit structure, and further involve another problem that the frequency of the clock for IC driving causes noise.
The above-described problems of the prior art techniques are described in, for example, "The Practical Power Source Circuit Design Hand Book", Jiro Togawa, CQ Publication, "The Practical Electronic Circuit Hand Book 4" CQ Publication and "The Transistor Technology Special No. 28" CQ Publication, etc.